Rake receiver

ABSTRACT

The invention relates to a Rake receiver of a CDMA system using IRC. The Rake receiver comprises at least two antenna branches, at least one Rake finger, and a delay estimator. The delay estimator comprises a despreader and an allocator for selecting at least one delay, and allocating a Rake finger for processing the signal component found by informing the Rake finger of the delay found. The delay estimator further comprises: a channel estimator, an interference estimator for generating an interference signal, a weighting coefficient part for providing each antenna branch with weighting coefficients maximizing the Signal-to-Interference-and-Noise Ratio, a multiplier for multiplying the pilot part by a weighting coefficient, and an antenna branch summer for combining the despread pilot parts, received via the separate antenna branches and multiplied by the weighting coefficient, to one combined pilot signal, on which combined pilot signal the selection is based in the allocator.

FIELD OF THE INVENTION

[0001] The invention relates to a Rake receiver of a radio system usinga Code Division Multiple Access (CDMA) method.

DESCRIPTION OF THE BACKGROUND ART

[0002] In radio systems, diversity methods of different kinds are usedfor increasing the coverage area and/or capacity of the system. As tothis publication, space diversity, i.e. antenna diversity, polarizationdiversity and multipath diversity are of interest. Space diversityindicates that antennas are positioned sufficiently far from each otherto achieve a sufficient decorrelation between signals received via theseparate antennas. An interesting kind of polarization diversity isimplicit polarization, i.e. a signal is sent on one polarization level,but received by cross-polarized antennas. Multipath diversity refers todiversity created by multipath propagated signal components, thisdiversity being usable in a system, such as a CDMA system, in which thebandwidth of a signal is much wider than the coherent bandwidth of achannel.

[0003] In a CDMA system, a Rake receiver is used for separatingmultipath propagated signal components at reception. In general, thesignal components must then be separated from each other by at least onechip of a spreading code used. The Rake receiver comprises Rake fingersand, in each of these fingers, despreading and diversity combinationtake place. In addition, the receiver comprises a delay estimator havinga matched filter for each antenna branch and an allocation block for theRake fingers. In the matched filter, a signal, received by a spreadingcode used for signal spreading, is correlated by different delays, thetiming of the spreading code then being changed for instance in steps ofone chip. When the correlation is high, a multipath propagated signalcomponent is found and it can then be received at the delay found.

[0004] On the radio path, the signal will include not only the desiredsignal but also noise and interference caused by other users or systems.In systems utilizing diversity, the influence of noise and interferencecan be decreased for instance by the Maximal Ratio Combining (MRC)method, in which signals received via separate antennas are weighted inproportion to the signal power in the separate antenna branches.However, this method presupposes that the interference of each antennais independent. This presupposition is not always true in actualcellular radio networks, but it is conceivable that the sameinterference is present at each antenna.

[0005] There is no such restriction on the Interference RejectionCombining (IRC) method. However, the method has been used only insystems utilizing the Time Division Multiple Access (TDMA) method, thesesystems often being incapable of separating multipath propagated signalcomponents. Herein, an IRC method refers to adaptive beam formation(optimal combination of signals), by which signal power is maximized inproportion to the power of interference and noise, i.e. theSignal-to-Interference-and-Noise Ratio (SINR) is maximized. Now we shallconcentrate on the code acquisition block, or delay estimator, of thereceiver. It consists of L matched filters and an allocator for Rakefinger allocation. The task of the matched filters is to match thespread and scrambled pilot sequence to the complex conjugated antennasignal in order to resolve the delays of the channel impulse responsetaps. In the Rake finger allocation, the temporal Rake fingers areallocated for the different multipath components of the received signal.

[0006] The matched filters can also be implemented as a bank of parallelcorrelators which carry out the correlation function of the complexconjugated spreading sequence. Each correlator performs despreadingprocedure which, mathematically, is the calculation of thecross-correlation function between the received signal and the cophasalcomplex conjugated spreading sequence.

[0007] The outputs of the correlators are used for allocating the Rakefingers to demodulate the strongest muitipath components of the receivedsignal. The current method of Rake finger allocation is based on theenergy of despread pilot symbols from L antennas. The outputs ofdespreaders are summed up at each code phase and N temporal Rake fingersare allocated according to the strongest energy of the sum signal. Inthe WCDMA (Wideband CDMA) concept the delay is estimated from thededicated physical control channel (DPCCH). The result is averaged overseveral time slots to get improved estimates for the Rake fingerallocation process.

[0008] The current Rake finger allocation is optimal in a spatiallywhite interference scenario, in which the interference sources areevenly distributed in the angular domain. In a spatially colouredinterference field, a high-powered interference source can decrease theperformance of the receiver because the Rake fingers are allocated tothe wrong chip delays.

BRIEF DESCRIPTION OF THE INVENTION

[0009] The present invention seeks to provide an improved Rake receiver.According to an aspect of the present invention, there is provided aRake receiver as specified in claim 1. The preferred embodiments of theinvention are claimed in the dependent claims.

[0010] The presented optimum combining scheme is capable of placingnulls towards interfering signals. Owing to this, the interference canbe suppressed in Rake finger allocation. The number of the wrong Rakefinger allocations can therefore be decreased, which improves theperformance of the receiver. The receiver can also better track thechanging interference field, and the spatial properties of interferencefield are taken into account in delay estimation.

LIST OF THE DRAWINGS

[0011] Embodiments of the present invention are described below, by wayof example only, with reference to the attached drawings, in which

[0012]FIGS. 1A and 1B illustrate a mobile telephone system;

[0013]FIG. 2A shows a transmitter and a receiver of a mobile telephonesystem;

[0014]FIG. 2B illustrates spreading and modulation in a transmitter;

[0015]FIG. 2C illustrates a combined descrambling, decoding anddemodulation block of the receiver of FIG. 2A;

[0016]FIG. 2D illustrates an embodiment of the delay estimator;

[0017]FIG. 2E illustrates another embodiment of the delay estimator;

[0018]FIG. 3 illustrates channels of a mobile telephone systempositioned in a frame;

[0019]FIG. 4 illustrates the structure of user equipment in a simplifiedmanner.

DESCRIPTION OF THE EMBODIMENTS

[0020] In the following examples, embodiments of the invention aredescribed in the Universal Mobile Telephone System (UMTS) withoutrestricting the invention to it.

[0021] The structure of a universal mobile telephone system is explainedreferring to FIGS. 1A and 1B. FIG. 1B comprises only the blocks that areessential for the description of the invention, but it is obvious to oneskilled in the art that a conventional mobile telephone system alsocomprises other functions and structures, which need not be explainedhere in more detail. The main parts of a mobile telephone system are aCore Network CN, a UMTS terrestrial radio access network UTRAN and UserEquipment UE. The interface between the CN and the UTRAN is called Iuand the air interface between the UTRAN and the UE is called Uu.

[0022] The UTRAN comprises Radio Network Subsystems RNS. The interfacebetween the RNSs is called lur. An RNS comprises a Radio NetworkController RNC and one or more nodes B. The interface between the RNCand B is called lub. The coverage area of node B, i.e. a cell, isdenoted by C in FIG. 1B.

[0023] The description in FIG. 1A is very abstract, and it is thereforeclarified in FIG. 1B, which shows the parts of the GSM system thatapproximately correspond to the parts of the UMTS. It should be notedthat the mapping presented is not in any way binding, but indicative,because the responsibilities and functions of the various parts of theUMTS are still under development.

[0024] In accordance with FIG. 1B, a circuit-switched connection can beestablished from the user equipment UE to a telephone 136 connected to aPublic Switched Telephone Network (PSTN) 134. The user equipment UE canbe for instance a fixed terminal, a terminal positioned in a vehicle ora portable terminal. The radio network infrastructure UTRAN comprisesradio network subsystems RNS, i.e. base station systems. The radionetwork subsystem RNS comprises a radio network controller RNC, i.e. abase station controller, and at least one node B, i.e. a base station,controlled by that controller.

[0025] A base station B comprises a multiplexer 114, transceivers 116and a control unit 118 controlling the operation of the transceivers 116and the multiplexer 114. Traffic and control channels used by aplurality of transceivers 116 are positioned on a transmission link lubby the multiplexer 114.

[0026] From the transceivers 116 of the base station B, there is aconnection to an antenna unit 120 implementing a bidirectional radioconnection Uu to the user equipment UE. The structure of the frames tobe transmitted on the bidirectional radio connection Uu is accuratelydefined.

[0027] The base station controller RNS comprises a switching network 110and a control unit 112. The switching network 110 is used for connectingspeech and data and for combining signalling circuits. The base stationsystem, comprising the base station B and the base station controllerRNC, additionally comprises a transcoder 108. The division of tasksbetween the base station controller RNC and the base station B and thephysical structure thereof may vary according to the implementation. Thebase station B typically attends to the implementation of the radio pathin the above-described manner. The base station controller RNC typicallycontrols things as follows: radio resources, handover between cells,power control, timing and synchronization, paging of user equipment.

[0028] The transcoder 108 is generally situated as close to a mobilephone exchange 106 as possible, because speech can then be transmittedin the form of a mobile phone system between the transcoder 108 and thebase station controller RNC, thus by saving transmission capacity. Thetranscoder 108 converts the various digital speech coding forms betweenthe public switched telephone network and the radio telephone networkinto a compatible format, for instance from a 64 kbit/s format of afixed network to another (for instance 13 kbit/s) format of the cellularradio network and vice versa. The devices required are not describedhere any further, but it can be stated that no other data than speech isconverted by the transcoder 108. The control unit 112 performs callcontrol and mobility management, collects statistic data and performssignalling.

[0029] A core network CN consists of an infrastructure belonging to amobile telephone system outside the UTRAN. Of the devices of the corenetwork CN, FIG. 1B illustrates the mobile phone exchange 106 and agateway mobile phone exchange 104 which attends to the connections ofthe mobile phone system to the outside world, here to the publicswitched telephone network 102.

[0030]FIG. 4 shows an example of the structure of the user equipment UE.The substantial parts of the user equipment UE are: an interface 404 foran antenna 402 of the user equipment, a transceiver 406, a control part410 of the user equipment and an interface 412 for a battery 414. A userinterface generally comprises a display 400, a keyboard 408, amicrophone 416 and a loudspeaker 418. The user equipment may be forinstance a portable mobile phone, a phone to be positioned in a car, aterminal of wireless local loop or data transmission equipmentintegrated into a computer.

[0031] The system can also employ packet-switched transmissionequipment, such as the GPRS (General Packet Radio Service). The GPRS(General Packet Radio Service) is a service in which air interfacecapacity not used in circuit-switching is employed for packettransmission. As the GPRS is a GSM-based emerging service, no details onthe adaptation thereof to the UMTS will be given.

[0032] As FIG. 1B shows, the switching field 110 can perform switching(depicted by black spots) to a public switched telephone network (PSTN)134 through the mobile services switching centre 106 and to a packettransmission network 142. A typical terminal 136 in the public switchedtelephone network 134 is an ordinary or an ISDN (Integrated ServicesDigital Network) phone.

[0033] The connection between the packet transmission network 142 andthe switching field 110 is established by a support node (SGSN=ServingGPRS Support Node) 140. The aim of the support node 140 is to transferpackets between the base station system and a gateway node (GGSN=GatewayGPRS Support Node) 144, and to keep record of the location of thesubscriber terminal UE within its area.

[0034] The gateway node 144 connects the packet transmission network 142and a public packet transmission network 146. An Internet protocol or anX.25 protocol can be used at the interface. By encapsulation, thegateway node 144 hides the internal structure of the packet transmissionnetwork 142 from the public packet transmission network 146, so for thepublic packet transmission network 146 the packet transmission network142 resembles a sub-network, the public packet transmission networkbeing able to address packets to the subscriber terminal UE placedtherein and to receive packets therefrom.

[0035] The packet transmission network 142 is typically a privatenetwork which uses an Internet protocol carrying signalling and userdata. As regards the architecture and protocols below the Internetprotocol layer, the structure of the network 142 may varyoperator-specifically.

[0036] The public packet transmission network 146 may be for example aglobal Internet to which a terminal 148, for example a server computer,with a connection thereto wants to transfer packets to the subscriberterminal UE.

[0037]FIG. 2A illustrates the function of a pair of radio transceivers.A radio transmitter may be located in a node B or in the user equipmentUE and a radio receiver in the user equipment UE or in the node B.

[0038] The upper part of FIG. 2A shows the essential functions of aradio transmitter. Various services to be located on a physical channelare for instance speech, data, moving or stopped video image and systemcontrol channels. The figure illustrates a control channel and dataprocessing. The various services require various source coding means;speech, for instance, requires a speech codec. However, for the sake ofclarity, the source coding means are not shown in FIG. 2A.

[0039] Pilot bits used by the receiver for channel estimation are alsolocated on the control channel 214. User data 200 is located on the datachannel.

[0040] The various channels are then channel-coded in various ways inblocks 202A and 202B. Channel coding comprises for instance differentblock codes, an example of them being Cyclic Redundancy Check (CRC). Inaddition, convolution coding and its various modifications, such aspunctured convolution coding or turbo coding, are typically used. Saidpilot bits are not channel-coded, however, because the intention is tofind out the signal distortions caused by the channel.

[0041] After the various channels have been channel-coded, they areinterleaved in an interleaver 204A, 204B. The aim of interleaving is tofacilitate error correction. At interleaving, the bits of the variousservices are scrambled together in a predetermined way, whereby aninstantaneous fading on the radio path alone does not necessarily makethe information transmitted unfit for identification. Subsequently, theinterleaved bits are spread by a spreading code in blocks 206A, 206B.The chips obtained are then scrambled by a scrambling code and modulatedin block 208, the operation of which is described in more detail in FIG.2B. In this way, the separate signals are combined in block 208 to betransmitted via the same transmitter.

[0042] Finally, the combined signal is brought to radio frequency parts210, which may comprise different power amplifiers and bandwidthrestricting filters. Regulation of a closed loop used for transmissionpower control generally controls a transmission power control amplifierlocated in this block. An analog radio signal is then sent via theantenna 202 to the radio path Uu.

[0043] The lower part of FIG. 2A illustrates the essential functions ofa radio receiver. The radio receiver is typically a Rake receiver. Ananalog radio frequency signal is received from the radio path Uu by anantenna 232. The signal is brought to radio frequency parts 230comprising a filter, which suppresses frequencies outside the desiredfrequency band.

[0044] Subsequently, the signal is converted in block 228 to anintermediate frequency or directly to baseband, in which form the signalis sampled and quantized. Because the signal is a multipath propagatedsignal, the intention is to combine the signal components propagatedalong different paths in block 228, the block comprising the actual Rakefingers of the receiver according to the prior art technique. Block 228is described in more detail in FIG. 2C.

[0045] The physical channel obtained is then deinterleaved in adeinterleaver 226. Subsequently, the deinterleaved physical channel isdivided into data streams of different channels in a demultiplexer 224.Each channel is brought to a separate channel decoding block 222A, 222B,where the channel coding used for a transmission, such as block codingand convolution coding, is decoded. Convolution coding is decodedpreferably by a Viterbi decoder. Each transmitted channel 220A, 220B canthen be brought to be further processed, as needed, for instance data220 is brought to a computer 122 connected to the user equipment UE. Thecontrol channels of the system are brought to a control part 236 of theradio receiver.

[0046]FIG. 2B illustrates in more detail how a channel is spread by aspreading code and modulated. To the left in the figure, a bit stream ofthe channel arrives at block SIP, where each two-bit sequence isconverted from series form to parallel form, which means that one bit isbrought to branch I of the signal and the other one to branch Q of thesignal. Subsequently, the signal branches I and Q are multiplied by aspreading code c_(ch), whereby relatively narrow-band information isspread to a wide frequency band. Each branch can have the same or adifferent spreading code. Each connection Uu has a separate spreadingcode or separate spreading codes, by which the receiver identifies thetransmissions intended for it. Then the signal is scrambled bymultiplication with a scrambling code c_(I scramb)+j c_(Q scramb), whichis separate for each transmitter. The pulse form of the obtained signalis filtered by filters p(t). Finally, the signal is modulated to a radiofrequency carrier by multiplying its separate branches shifted from eachother by 90 degrees, the branches thus obtained are combined to onecarrier ready to be sent to the radio path Uu, excluding possiblefilterings and power amplifications. The modulation mode described isQuadrature Phase Shift Keying QPSK.

[0047] Instead of the described I/Q multiplexing, time multiplexing canalso be used, where data and control channels are positionedsequentially on the time axis. However, the time difference between thechannels is then so small that an interference estimated from thecontrol channel can be assumed to be the same also on the data channel.

[0048] Maximally, 256 different mutually orthogonal spreading codes cantypically be -used simultaneously. For instance, if the UMTS uses a fivemegahertz carrier at the speed of 4.096 megachips per second in thedownlink direction, the spreading factor 256 corresponds to thetransmission speed of 32 kbit/s, and respectively, the highest practicaltransmission speed is achieved by spreading factor four, whereby thedata transmission speed is 2048 kbit/s. Accordingly, the transmissionspeed on the channel varies stepwise from 32, 64,128, 256, 512,1024 to2048 kbit/s, the spreading factor being 256, 128, 64, 32, 16, 8 and 4,respectively. The data transmission speed at the user's disposal dependson the channel coding used. For instance, if ⅓ convolution coding isused, the user's data transmission speed is about one third of the datatransmission speed of the channel. The spreading factor informs thelength of the spreading code. For instance, the spreading codecorresponding to the spreading factor one is (1). The spreading factortwo has two mutually orthogonal spreading codes (1,1) and (1,−1).Further, the spreading factor four has four mutually orthogonalspreading codes: below an upper level spreading code (1,1), there arespreading codes (1,1,1,1) and (1,1,−1,−1), and below another upper levelspreading code (1,−1), there are spreading codes (1,−1,1,−1) and(1,−1,−1,1). The formation of spreading codes is continued in this waywhen propagating to lower levels of a code tree. The spreading codes ofa given level are always mutually orthogonal. Likewise, a spreading codeof a given level is orthogonal to all the spreading codes of anotherspreading code of the same level, which are derived from that otherspreading code to next levels.

[0049] In transmission, one symbol is multiplied by a spreading code,whereby the data spreads over the frequency band to be used. Forinstance, when the spreading code 256 is used, one symbol is representedby 256 chips. Respectively, when the spreading code 16 is used, onesymbol is represented by 16 chips.

[0050]FIG. 3 shows an example of which kind of frame structure can beused on a physical channel. Frames 340A, 340B, 340C, 340D are numberedconsecutively from one to 72 and they form a 720 milliseconds longsuperframe. The length of one frame 340C is 10 milliseconds. The frame340C is divided into sixteen slots 330A, 330B, 330C, 330D. The length ofone slot 330C is 0.625 milliseconds. One slot 330C corresponds typicallyto one power control period, during which the power is controlled forinstance by one decibel upwards or downwards.

[0051] The physical channels are divided into two different types:Dedicated Physical Data Channels (DPDCH) 310 and Dedicated PhysicalControl Channels (DPCCH) 312. The dedicated physical data channels 310are used for transporting data 306 generated in layer two of OpenSystems Interconnection (OSI) and above it, i.e. dedicated trafficchannels, mainly. The dedicated physical control channels 312 transportcontrol information generated in layer one of OSI. The controlinformation comprises: a pilot part, i.e. pilot bits, 300 to be utilizedfor channel estimation, Transmit Power Control (TPC) commands 302 and,optionally, a Transport Format Indicator (TFI) 304. The transport formatindicator 304 tells the receiver the transmission speed used at thatmoment by each dedicated physical data channel in the uplink direction.

[0052] As appears from FIG. 3, the dedicated physical data channels 310and the dedicated physical control channels 312 are time-multiplexedinto the same slot 330C in the downlink direction. In the uplinkdirection again, these channels are transmitted in parallel in such away that they are IQ-multiplexed (I=Inphase, Q=Quadrature) into eachframe 340C and transmitted by dual-channel Quadrature Phase Shift Keying(QPSK) modulation. When the intention is to transmit additionaldedicated physical data channels 310, they are code-multiplexed eitherinto branch I or Q of the first pair of channels.

[0053] Subsequently, FIG. 2C is examined, the figure illustrating inmore detail the combined descrambling, decoding and demodulating block228 of the receiver, shown in FIG. 2A. Descrambling is not described,however, because it is of no relevance to the invention. A desired radiosignal, sent to the radio path Uu, multipath propagates on anoccasionally fading channel 250. Further, additive zero mean whitegaussian noise 254 is combined with the signal. Moreover, interferingsignals, also multipath propagating on an occasionally fading channel252, are combined with the signal.

[0054] Consequently, a signal to be received from the radio path Uucontains, not only the desired signal, but also both noise andinterference. The signal is received by at least two separate antennabranches 232A, 232B. The branches 232A, 232B may form an antenna arrayto provide antenna gain, the separate antennas being relatively close toeach other, at a distance of half a wavelength, for instance. Anotherpossibility is that the branches 232A, 232B are diversity branches, theseparate antennas being relatively far from each other, at a distance of10 to 20 wavelengths, for instance. The diversity can be implemented asspace or polarization diversity.

[0055] The example of FIG. 2C illustrates the use of space diversity,the branches 232A, 232B being implemented as an adaptive antenna. Theadaptive antenna is implemented by antennas 232A, 232B positioned farenough from each other, via which antennas the multipath propagatedsignal is received.

[0056] The number of antennas may be L. The figure illustrates only twoantennas, the first antenna 232A and the Lth antenna 232B. The twopoints between the antennas represent the existing antennas, but are notdescribed for the sake of clarity. Generally, the number of antennasvaries between two and eight.

[0057] In accordance with the invention, signals received via theseparate antenna branches 232A, 232B are weighted in such a way that theinfluence of noise and interference can be minimized.

[0058] When diversity is used, the intention is to make the correlationbetween the branches as low as possible. Another way of implementingdiversity is to use polarization diversity, whereby a signal is receivedby cross-polarized antennas. In theory, also hybrids are possible, whichmeans that both space and polarization diversity may be usedsimultaneously. An example of a solution applicable to user equipment isa so-called patch antenna, which can be a plate of about a square inchin size, the plate having cross polarization planes. Another example isuser equipment positioned in a vehicle, where an implementation of spacediversity is also relatively easy.

[0059] A signal received from all L antenna branches 232A, 232B isbrought via radio frequency parts (not shown in FIG. 2C) to a delayestimator 260 connected to the antenna branch 232A, 232B. In the delayestimator 260, the delays of the best audible multipath propagatedsignal components are searched for. A Rake finger 270A, 270B isallocated for processing the found multipath propagated signalcomponents. The delay estimator 260 informs each Rake branch 270A, 270Bof the delay found.

[0060] The delay estimator 260 comprises a matched filter 262A, 262B foreach antenna branch 232A, 232B. Thus, the number of matched filters262A, 262B is also L. In the matched filter 262A, 262B, a predeterminednumber of parallel correlation calculations are performed for thereceived radio signal by different delays in order to estimate thedelays of the multipath propagated signal components. In correlationcalculation, the spread pilot part contained in the received radiosignal is despread by a known spreading code using a predetermineddelay.

[0061] On the basis of the calculated correlations, an allocator 264situated in the delay estimator selects at least one delay, by which amultipath propagated signal component is received. The allocatorallocates a Rake finger 270A, 270B for processing the signal componentfound by informing the Rake finger of the delay found. To perform theselection, the correlation results of each matched filter 262A, 262B aretypically combined in the allocator 264. If the correlation is high, adelay is found that represents the delay of the multipath propagatedsignal component of the radio signal coming to the antenna branch 232A,232B in question. In general, the strongest multipath components havethe same code phases at all antennas, which is due to the vicinity ofthe antennas and to the fact that radio signals propagate at the speedof light.

[0062] As was said earlier, another known method for Rake fingerallocation is based on the energy of despread pilot symbols from Lantennas. The outputs of despreaders are summed up at each code phaseand N temporal Rake fingers are allocated according to the strongestenergy of the sum signal.

[0063]FIG. 2D shows an embodiment of the delay estimator. One delayestimator 290A, 290B processes one multipath propagated signal componentby a given code delay. The delay estimators 290A, 290B are describedhere as different instances for the sake of clarity, but they can alsobe realized as one instance operating internally in parallel. The delayestimator 290A, 290B comprises a channel estimator 292, by which achannel impulse response of a multipath propagated signal component,included in a radio signal and found by means of a known pilot part,i.e. practically complex impulse response taps of the channel, isgenerated.

[0064] In addition, the delay estimator 290A, 290B comprises aninterference estimator 292, by which an interference signal, included ina radio signal of each antenna branch 232A, 232B and consisting ofinterference and noise, is generated. The interference signal can begenerated by any means known to a person skilled in the art. In anembodiment the interference estimator 292 generates an interferencesignal by subtracting from the received radio signal a desiredregenerated radio signal. In this embodiment the desired regeneratedradio signal is obtained by means of the known pilot part and theestimated impulse response of the channel. For improved performance alsodecision feedback of detected data bits included in the despreadmultipath propagated signal component can be employed in the estimationof the impulse response of the channel and the interference.

[0065] In another embodiment the interference estimator 292 generatesthe interference signal by using multi-user detection, whereby thesignals of other users form the interference signal. More information onmultiuser detection can be found in the following article: Verdu,Sergio: Adaptive Multiuser detection, published in IEEE ISSSTA '94Proceedings of the IEEE Third International Symposium on Spread SpectrumTechniques and Applications, ISBN 07803-1750-5, which is incorporatedherein by reference.

[0066] The delay estimator 290A, 290B, comprises a despreader 296A,296B, which is connected to each antenna branch 232A, 232B and despreadsthe spread pilot part included in the multipath propagated signalcomponent, by a known spreading code at a delay.

[0067] There are L despreaders for processing the pilot part, i.e. oneper each antenna branch 232A, 232B in each delay estimator 290A, 290B.In practice, when despreading, the pilot part of the signal component ismultiplied by a complex conjugate of the spreading code in the rightphase.

[0068] A weighting coefficient part 292 in the delay estimator 290A,290B forms weighting coefficients maximizing thesignal-to-interference-and-noise ratio (SINR) for each antenna branch232A, 232B. This can be made for instance by multiplying an inversematrix of a covariance matrix of an interference signal, consisting ofinterference and noise of the antenna branches 232A, 233B, by anestimated impulse response vector of the channel. The weightingcoefficients are complex.

[0069] The pilot part despread by the despreader 296A, 296B in eachantenna branch 232A, 232B is multiplied by the obtained weightingcoefficients by using a multiplier 294A, 294B located in the delayestimator 290A, 290B.

[0070] An antenna branch summer 298, positioned last in the delayestimator 290A, 290B, combines the despread pilot parts, received viathe separate antenna branches 232A, 232B and multiplied by a weightingcoefficient, to one pilot signal.

[0071] As a whole, the situation is such that the delay estimator 290A,290B allocates N Rake fingers 270A, 270B, for the best audible signalcomponents. The outputs of the despreaders of different antenna branchesare summed up at each code phase and N temporal Rake fingers areallocated according to the strongest energy of the sum signal.

[0072] On the basis of the energies of the formed pilot signals, anallocator 264 situated in the delay estimator selects at least onedelay, by which a multipath propagated signal component is received.Instead of energy values power values or calculated correlation valuesmay also be used. Pilot signals having the highest energies areselected. The allocator 264 allocates a Rake finger 270A, 270B forprocessing the signal component found by informing the Rake finger ofthe delay found. The number N may vary depending on the circumstances,or a threshold value may be set for the level of the multipathpropagated signal component. Consequently, the search for timing is adynamic process, and so is the allocation of the Rake fingers 270A, 270Bto be combined.

[0073] In practice, a predetermined number of Rake fingers 270A, 270B,are allocated and/or a number required for delays exceeding apredetermined threshold value at correlation calculation. Generally, alimiting factor will be the maximum number of the Rake fingers 270A,270B used. In this example, the number of allocated Rake fingers 270A,270B is indicated by the letter N. The number of signal componentsdepends on radio conditions and, for instance, on terrain shape andbuildings causing reflections. In most cases, the smallest delay bywhich multipath propagated signal components are searched for is onechip. The frequency of Rake finger allocation can be variable. It can beperformed for each slot or each frame, for example.

[0074] The functioning of the delay estimator 290A can be improved bythree separate filter structures. These three solutions can be usedalone or combined in any way. The impulse response of the channelgenerated by the channel estimator 292 is averaged coherently by a firstfilter structure connected to the channel estimator 292. The betterchannel estimate thus obtained also makes the weighting coefficientsmore reliable. The despread pilot part multiplied by the weightingcoefficient is non-coherently filtered by a second filter structureconnected between the multiplier 294A, 294B and the antenna branchsummer 298 in each antenna branch 232A, 232B. This improves the resultobtained in each antenna branch 232A, 232B. The combined pilot signal isaveraged non-coherently by a third filter structure connected betweenthe antenna branch summer 298 and the allocator 264.

[0075] One Rake finger 270A, 270B processes one multipath propagatedsignal component by a given code delay. The Rake finger 270A, 270Bcomprises a channel estimator 272, by which a channel impulse responseof a multipath propagated signal component, included in a radio signaland found by means of a known pilot part, i.e. practically compleximpulse response taps of the channel, is generated.

[0076] In addition, the Rake finger 270A, 270B comprises an interferenceestimator 272, by which an interference signal, included in a radiosignal of each antenna branch 232A, 232B and consisting of interferenceand noise, is generated by subtracting from the received radio signal adesired regenerated radio signal. The desired regenerated radio signalis obtained by means of the known pilot part included in the radiosignal and by means of the estimated impulse response of the channel.

[0077] The areas drawn with broken lines in FIG. 2C illustrate theprocessing of the pilot part 274A included in the radio signal and theprocessing of the data part 274B included in the radio signal.

[0078] The Rake finger 270A, 270B, comprises a despreader 276A, 276B,connected to each antenna branch 232A, 232B and despreading the spreadpilot part 274A included in the multipath propagated signal component,by using a known spreading code by a delay informed by the delayestimator 260.

[0079] Correspondingly, the Rake finger 270A, 270B comprises adespreader 276C, 276D, which is connected to each antenna branch 232A,232B and despreads the spread data part 274B included in the multipathpropagated signal component, by a known spreading code by a delayinformed by the delay estimator 260. There are L despreaders forprocessing both the data part and the pilot part, i.e. two per eachantenna branch 232A, 232B in each Rake finger 270A, 270B. In practice,when despreading, the data part or the pilot part of the signalcomponent is multiplied by a complex conjugate of the spreading code inthe right phase.

[0080] As a whole, the situation is such that the delay estimator 260allocates N Rake fingers 270A, 270B, for the best audible signalcomponents. In each Rake finger 270A, 270B, L antenna branches 232A,232B are processed. Both the pilot part of the radio signal and the datapart of the radio signal are processed separately. The number N may varydepending on the circumstances, or a threshold value may be set for thelevel of the multipath propagated signal component. If this thresholdvalue is exceeded, said Rake finger 270A, 270B is notified and thereception continues. Consequently, the search for timing is a dynamicprocess, and so is the allocation of the Rake fingers 270A, 270B to becombined.

[0081] A weighting coefficient part 272 in the Rake finger 270A, 270Bforms weighting coefficients which maximize thesignal-to-interference-and-noise ratio (SINR) for each antenna branch232A, 232B. This can be carried out for instance by multiplying aninverse matrix of a covariance matrix of an interference signal,consisting of interference and noise of the antenna branches 232A, 233B,by an estimated impulse response of the channel. The weightingcoefficients are complex.

[0082] The pilot part 274A despread by the despreader 276A, 276B in eachantenna branch 232A, 232B is multiplied by the obtained weightingcoefficients by a multiplier 284A, 284B located in the Rake finger 270A,270B.

[0083] Correspondingly, the data part 274B despread by the despreader276C, 276D in each antenna branch 232A, 232B is multiplied by theobtained weighting coefficients by a multiplier 284C, 284D. Accordingly,the signal components including the pilot part and the signal componentsincluding the data part are multiplied by the same weightingcoefficients separately.

[0084] An antenna branch summer 278A, positioned last in the Rake finger270A, 270B, combines the despread pilot parts 274A, received via theseparate antenna branches 232A, 232B and multiplied by a weightingcoefficient, to one pilot signal.

[0085] Correspondingly, an antenna branch summer 278B combines thedespread data parts 274B, received via the separate antenna branches232A, 232B and multiplied by a weighting coefficient, to one datasignal.

[0086] The Rake receiver additionally comprises a Rake finger summer280B combining the data signals of the Rake fingers 270A, 270Bfunctioning by different delays to a sum data signal representing thereceived bits. The data bits are then brought according to FIG. 2A fromblock 228 to block 226 to be deinterleaved.

[0087] The receiver presented is suitable for use both at a base stationand at user equipment. This means that both I/Q multiplexing and timemultiplexing of data channel and control channel are possible.

[0088] Between the antenna branch summer 278A, 278B and the Rake fingersummer 280A, 280B, there may be a real part 278A, 278B, removing fromthe combined signal of each antenna branch its imaginary part, becausethe imaginary part is an error term generated during channel estimation.

[0089] In a preferred embodiment, the Rake receiver comprises a Rakefinger summer 280A combining the pilot signals of the Rake fingers 270A,270B, functioning by different delays, to a sum pilot signalrepresenting the received pilot bits. This sum pilot signal can bebrought to an estimator 282 for the signal-to-inference ratio,estimating the signal-to-interference ratio of said channel. The powercontrol of a closed loop can be controlled by the obtainedsignal-to-interference ratio of said channel. This is illustrated inblock 282 of FIG. 2C by the text TPC (Transmission Power Control). Theinvention is implemented preferably by software, at least part of thefunctions included in block 228 being changed to software to beperformed by a processor. However, the delay estimator 260, 290Arequiring a high calculation capacity is preferably implemented as anApplication Specific Integrated Circuit (ASIC). The other functionsincluded in block 228 can also be implemented by device solutionsoffering the needed functionality, such as an ASIC or a discrete logic.

[0090] A method of calculating weighting coefficients maximizing theSINR is presented next, assuming that the impulse response h of thechannel and the covariance matrix RUU of interference and noise areknown. The method can be used both in the Rake fingers 270A, 270B and inthe delay estimator 290A. Subsequently, a method of estimating h andR_(uu) by means of known pilot bits included in a signal is presented.The presentation is a complex baseband signal model on symbol level forprocessing the signal. In the presentation, the bold face termsillustrate a vertical vector or a matrix. Let us assume. that Nmultipath propagated Signals Of Interest (SOI) are found on time axis bymatched filters, and each signal component is received via L separateantennas. The L complex channel taps of the Nth multipath propagatedsignal component are indicated by vectors h_(n) having a length L. Theadditive Multi Access Interference (MAI) caused by other users,multipath self-interference and noise are indicated by a vector u_(n),which is modelled as an L-variate complex Gaussian distributed processwith spatial possibly coloured covariance R_(uu,n)=E[u_(n)u_(n) ^(H)].The signal received from the L antennas is indicated by a vector r_(n).An information symbol of the Mth user out of an alphabet of size M isindicated by the term s_(m).

[0091] The Gaussian assumption for the despread MAI is valid for a greatnumber of spreading factors having different lengths.

[0092] Subsequently, each symbol period is discretized into K samples,whereby the vector r_(n) can be presented in the form:

r _(n) [k]=h _(n) s _(m) [k]+u _(n) [k], k=1, . . . ,K  (1)

[0093] By stacking each of the N vectors to vectors having a length LN,a more compact notation is obtained:

r[k]=hs _(m) [k]+u[k], k=1, . . . ,K  (2)

[0094] The Gaussian distributed interference variables u_(n)[k] and u[k]are mutually uncorrelated across sampling instants and also across thedifferent multipath propagated components of SOI. Then: $\begin{matrix}{{R_{uu}\lbrack k\rbrack} = {{E\left\lbrack {{u\lbrack k\rbrack}{u^{H}\lbrack k\rbrack}} \right\rbrack} = {{diag}\quad \left( {{R_{{uu},1}\lbrack k\rbrack},\ldots \quad,{R_{{uu},N}\lbrack k\rbrack}} \right)}}} & (3)\end{matrix}$

[0095] Assuming that the symbols s_(m) are equi-probable and the channelparameters h and the covariance matrix R_(uu)[k] of interference andnoise are both known, the optimal demodulation involves the maximizationof the log likelihood function (|·| denotes determinant):$\begin{matrix}\begin{matrix}{{L\left( {r,s_{m}} \right)} = {\ln \left( {\prod\limits_{k = 1}^{K}{\frac{1}{\pi^{LN}{{R_{uu}\lbrack k\rbrack}}}\exp \left\{ {{- {u\lbrack k\rbrack}}{R_{uu}^{- 1}\lbrack k\rbrack}{u^{H}\lbrack k\rbrack}} \right\}}} \right)}} \\{= {{- {\sum\limits_{\lambda = 1}^{K}{\left( {{r\lbrack k\rbrack} - {{hs}_{m}\lbrack k\rbrack}} \right)^{H}{R_{uu}^{- 1}\lbrack k\rbrack}\quad \left( {{r\lbrack k\rbrack} - {{hs}_{m}\lbrack k\rbrack}} \right)}}} + {const}_{1}}}\end{matrix} & (4)\end{matrix}$

[0096] Assuming that the symbols have the same energy, formula 4 can bedeveloped into the form: $\begin{matrix}\begin{matrix}{{L\left( {r,s_{m}} \right)} = {{\sum\limits_{k = 1}^{K}{2{Re}\quad \left\{ {{r^{H}\lbrack k\rbrack}\quad {R_{uu}^{- 1}\lbrack k\rbrack}\quad h\quad {s_{m}\lbrack k\rbrack}} \right\}}} + {const}_{2}}} \\{= {{2{Re}\quad \left\{ {\sum\limits_{k = 1}^{K}{\left( {\sum\limits_{n = 1}^{N}{{w_{n}^{H}\lbrack k\rbrack}{r_{n}\lbrack k\rbrack}}} \right){s_{m}^{*}\lbrack k\rbrack}}} \right\}} + {const}_{2}}} \\{= {2{Re}\quad \left\{ {s_{m}^{H} - t} \right\}}}\end{matrix} & (5)\end{matrix}$

[0097] whereby the N weighting coefficients minimizing the interferenceare w_(n)[k]=R_(uu,n) ⁻¹[k]h_(n), and the vectors s_(m) and t have alength K with elements s_(m)[k], respectively$\sum\limits_{n = 1}^{N}{{w_{n}^{H}\lbrack k\rbrack}\quad {{r_{n}\lbrack k\rbrack}.}}$

[0098] Accordingly, the IRC Rake receiver presented earlier can bedecomposed into N temporal Rake fingers, each of which performs spatialIRC on the L antenna inputs by using weighting coefficientsw_(n)[k]=R_(uu,n) ⁻¹[k]h_(n). The outputs of the Rake fingers aresummed, i.e, combined, and a correlation detector is applied todetermine for the symbol sm a value enabling the largest symbolcorrelation metric.

[0099] If the multipath self-interference of SOI can be neglected, forinstance when the processing gain is large enough, the R_(uu,n) isessentially the same in all N fingers, which means that it needs to beestimated and inverted only once. When the interference covariancematrix is spatially white, i.e. R_(uu,n)=Id, IRC becomes MRC, becausew_(n)[k]=h_(n). Direct Matrix Inversion (DMI) of the matrix R_(uu,n) canbe avoided, if recursive algorithms, such as Least Mean Square (LMS) orRecursive Least Square (RLS), are used. Accordingly, the receiver can beconstructed in such a way that the interference elimination method canbe changed according to the circumstances between the MRC and IRC. Whendata transmission speeds are high, the interference is coloured, andtherefore, IRC is used, and, respectively, MRC is used at low datatransmission speeds. In principle, MRC is only one special case of IRC,which means that the method to be used can always be IRC.

[0100] Assuming that h and R_(uu) are not known, an unstructured MaximumLikelihood ML channel estimation of vector h and an estimation of thecovariance matrix R_(uu) utilizing the performed channel estimation arepresented next. As stated earlier, I/Q multiplexing is used in theuplink direction, the data channel being multiplexed to the branch I andthe control channel to the branch Q. The control channel also comprisesa previously known pilot part. Both channels can be separated from eachother by despreading with orthogonal spreading codes. The symbol-levelsignal model is obtained from equation 1, by writing it separately foreach part, I and Q, using BPSK symbols s_(m)ε{−1,1}. It is furtherassumed that the index k now refers to the bit index of the symbolsequence. K bits of DPCCH are collected into one slot.

[0101] Previously, the channel parameters h and the interferencecovariance R_(uu) were assumed to be known. Now, it is assumed that no apriori information on either spatial structure is available, which meansthat the optimal channel estimates are created on the maximum likelihoodprinciple. The vector r[k], k=1, . . . ,K and the pilot bits s_(p)[k] ofthe DPCCH within one slot are used, by which ML estimates [ĥ,{circumflex over (R)}_(uu) are generated, being the joint minimizers ofthe log likelihood function: $\begin{matrix}\begin{matrix}{{L\left( {r,h,R_{uu}} \right)} = \quad {\ln \quad \left( {\prod\limits_{k = 1}^{K}{\frac{1}{\pi^{LN}{R_{uu}\lbrack k\rbrack}}\exp \left\{ {- \left( {{r\lbrack k\rbrack} -} \right.} \right.}} \right.}} \\\left. \left. {\left. \quad {{hs}_{p}\lbrack k\rbrack} \right)^{H}{R_{uu}^{- 1}\lbrack k\rbrack}\left( {{r\lbrack k\rbrack} - {{hs}_{p}\lbrack k\rbrack}} \right)} \right\} \right) \\{{{{{= \quad {- \ln}}}R_{uu}^{- 1}}} - {{trace}\left\{ {R_{uu}\frac{1}{K}{\sum\limits_{k = 1}^{K}\left( {{r\lbrack k\rbrack} - {{hs}_{p}\lbrack k\rbrack}} \right)}} \right\}} + {const}_{1}}\end{matrix} & (6)\end{matrix}$

[0102] This ML estimating problem is separable. When ML is given theestimate ĥ, the vector will be {circumflex over (R)}_(uu):$\begin{matrix}{{\hat{R}}_{uu} = {\frac{1}{K}{\sum\limits_{k = 1}^{K}{\left( {{r\lbrack k\rbrack} - {\hat{h}{s_{p}\lbrack k\rbrack}}} \right)\quad \left( {{r\lbrack k\rbrack} - {\hat{h}{s_{p}\lbrack k\rbrack}}} \right)^{H}}}}} & (7)\end{matrix}$

[0103] and the ML estimate ĥ is obtained as the minimizer of the costfunction (|·| denotes determinant): $\begin{matrix}\begin{matrix}{F = {{\frac{1}{K}{\sum\limits_{k = 1}^{K}{\left( {{r\lbrack k\rbrack} - {{hs}_{p}\lbrack k\rbrack}} \right)\quad \left( {{r\lbrack k\rbrack} - {{hs}_{p}\lbrack k\rbrack}} \right)^{H}}}}}} \\{= {{{\left( {h - {\hat{r}}_{sr}^{H}} \right)\quad \left( {h - {\hat{r}}_{sr}^{H}} \right)^{H}} + {\hat{R}}_{rr} - {{\hat{r}}_{sr}^{H}{\hat{r}}_{sr}}}}}\end{matrix} & (8)\end{matrix}$

[0104] F is minimized for the choice: $\begin{matrix}{h = {\hat{r}}_{sr}^{H}} & (9)\end{matrix}$

[0105] Instead of estimating R_(uu) from the despread signal, thewideband signal can be used for the covariance matrix estimation. Inthat approach we calculate R_(rr) instead of R_(uu) and use that term tosuppress the interference. In the R_(rr) estimation we have lots ofsamples and due to that the accuracy of the estimation can be increased.Also, in that approach, the covariance matrix needs to be calculated andinverted only once for all chip delay positions. So the computationalload can be decreased. The R_(uu) is the spatial correlation matrix ofthe interference plus noise and the R_(rr) is the spatial correlationmatrix of the signal plus interference plus noise. The R_(uu) approachis described in FIG. 2D, and the R_(rr) approach in FIG. 2E. FIG. 2E isotherwise the same as FIG. 2D but the interference estimator 286estimates the R_(rr) from the received wideband signal, and passes thisinformation to the weighting coefficient part 288.

[0106] A linear channel estimator based on pilot bits has been describedabove. It is obvious to one skilled in the art that known more developedchannel estimation methods, such as methods utilizing a data channel aswell, can be applied to the method of the invention.

[0107] In the radio system described, there may occur interferencecaused by the frequency band adjacent to the desired channel in somesituations, this interference being known as Adjacent Channel Power(ACP). The adjacent frequency band may be the WCDMA frequency bandadjacent to said operator, the WCDMA frequency band of another operatoror a frequency band of some other system, for instance the GSM system.The problem may cause blocking in the cell in the uplink direction. Forinstance, let us assume that a high efficiency GSM transmitter causesACP to a Rake receiver operating at a high data speed, i.e. at a lowspreading ratio, on a 5-MHz frequency band, for instance. The ACP (asinterference in general) must be above the noise level so that it can beeliminated. In accordance with the invention, an interference signalgenerated by the interference estimator 272 then comprises interferencecaused by the adjacent frequency band of the desired channel, i.e.adjacent channel power, the detrimental effect of which can thus beeliminated. A shrinking of the cell on account of ACP can thus beprevented.

[0108] Though the invention has been described above with reference tothe example of the attached drawings, it is clear that the invention isnot restricted to that, but can be modified in many ways within thescope of the inventive idea of the attached claims.

1. A Rake receiver comprising at least two antenna branches (232A, 232B)for receiving a radio signal, at least one Rake finger (270A, 270B)connected to the antenna branches (232A, 232B) for processing amultipath propagated signal component of the radio signal, and a delayestimator (290A) connected to the antenna branches (232A, 232B), thedelay estimator (290A) comprising: a despreader (296A, 296B) connectedto each antenna branch (232A, 232B) for despreading the pilot partincluded in the multipath propagated signal component by using a knownspreading code by a delay; and an allocator (264) for selecting at leastone delay, by which delay a multipath propagated signal component isreceived, and allocating a Rake finger (270A, 270B) for processing thefound signal component; characterized in that the delay estimator (290A)further comprises: a channel estimator (292) for generating an impulseresponse of the channel of the multipath propagated signal componentfound by means of a known pilot part included in the radio signal ofeach antenna branch (232A, 232B); an interference estimator (292) forgenerating an interference signal, included in the radio signal of eachantenna branch (232A, 232B) and consisting of interference and noise; aweighting coefficient part (292) for providing each antenna branch(232A, 232B) with weighting coefficients maximizing the signal-tointerference-and-noise ratio (SINR); a multiplier (294A, 294B) formultiplying the pilot part, despread by the despreader (296A, 296B) ineach antenna branch (232A, 232B), by a weighting coefficient; an antennabranch summer (298A) for combining the despread pilot parts, receivedvia the separate antenna branches (232A, 232B) and multiplied by theweighting coefficient, to one combined pilot signal, on which combinedpilot signal the selection is based in the allocator (264).
 2. A Rakereceiver according to claim 1 , characterized in that the interferenceestimator (292) generates one interference signal that is used for alldifferent delays.
 3. A Rake receiver according to claim 1 ,characterized in that the interference estimator (292) generatesinterference signal for each delay.
 4. A Rake receiver according toclaim 3 , characterized in that for each delay its own interferencesignal is used.
 5. A Rake receiver according to claim 3 , characterizedin that an average interference signal is calculated using theinterference signals of each delay, and this average interference signalis used for each delay.
 6. A Rake receiver according to any precedingclaim, characterized in that the interference estimator (292) uses as aninput the despread pilot part (274A) included in the multipathpropagated signal component by using a known spreading code by a delay.7. A Rake receiver according to any preceding claim 1 to 5,characterized in that the interference estimator (292) uses as an inputthe received radio signal.
 8. A Rake receiver according to any precedingclaim, characterized in that the interference estimator (292) generatesan interference signal by subtracting from the received radio signal adesired regenerated radio signal.
 9. A Rake receiver according to claim8 , characterized in that the desired regenerated radio signal isobtained by means of the known pilot part and the estimated impulseresponse of the channel.
 10. A Rake receiver according to claim 8 or 9 ,characterized in that the desired regenerated radio signal is obtainedby means of decision feedback of detected data bits included in thedespread multipath propagated signal component.
 11. A Rake receiveraccording to any preceding claim, characterized in that the interferenceestimator (292) generates an interference signal by using multi-userdetection, whereby the signals of other users form the interferencesignal.
 12. A Rake receiver according to any preceding claim,characterized in that the impulse response of the channel generated bythe channel estimator (292) is averaged coherently by a first filterstructure connected to the channel estimator (292).
 13. A Rake receiveraccording to any preceding claim, characterized in that the despreadpilot part multiplied by the weighting coefficient is non-coherentlyfiltered by a second filter structure connected between the multiplier(294A, 294B) and the antenna branch summer (298) in each antenna branch(232A, 232B).
 14. A Rake receiver according to any preceding claim,characterized in that the combined pilot signal is averagednon-coherently by a third filter structure connected between the antennabranch summer (298) and the allocator (264).
 15. A Rake receiveraccording to any preceding claim, characterized in that the despreader(296A, 296B) is replaced by a matched filter (262A, 262B) for eachantenna branch (232A, 232B) for performing a predetermined number ofparallel correlation calculations for the received radio signal bydifferent delays, whereby the pilot part included in the received radiosignal is despread in correlation calculation by a known spreading codeat a predetermined delay.
 16. A Rake receiver according to any precedingclaim, characterized in that, to provide antenna gain, the antennabranches (232A, 232B) form an antenna array, by which an antenna beam isformed in the desired direction by phasing separate antenna signals. 17.A Rake receiver according to any preceding claim, characterized in thatthe antenna branches (232A, 232B) are diversity branches.
 18. A Rakereceiver according to claim 17 , characterized in that the antennabranches (232A, 232B) are antennas implemented by space diversity.
 19. ARake receiver according to claim 17 , characterized in that the antennabranches (232A, 232B) are antennas implemented by polarizationdiversity.
 20. A Rake receiver according to any preceding claim,characterized in that the channel estimator (272) performs the channelestimation on the optimal Maximum Likelihood principle.
 21. A Rakereceiver according to any preceding claim, characterized in thatweighting coefficients maximizing the signal-to-interference-and-noiseratio are formed for each antenna branch (232A, 232B) by multiplying aninverse matrix of a covariance matrix, generated of an interferencesignal of the antenna branches (232A, 232B), by an estimated impulseresponse of the channel.
 22. A Rake receiver according to claim 21 ,characterized in that a channel estimate generated by the optimalMaximum Likelihood method is utilized for estimating the covariancematrix generated of interference and noise.
 23. A Rake receiveraccording to any preceding claim, characterized in that the interferencesignal generated by the interference estimator (272) comprisesinterference caused by the adjacent frequency band of the desiredchannel, i.e. adjacent channel power.